1. Field of the Invention
The present invention relates to a switching power supply circuit.
2. Description of the Related Art
As so-called soft-switching power supply of a resonant type, a current resonant type and a voltage resonant type are widely known. In a present situation, a current resonant converter of a half-bridge coupling system using two switching devices is in wide use because such a current resonant converter is easily put to practical use.
However, the characteristics of a high withstand voltage switching element, for example, are now being improved, and therefore the problem of withstand voltage in putting a voltage resonant converter to practical use is being cleared up. In addition, a voltage resonant converter formed as a single-ended type with one switching element is known to be advantageous as compared with a current resonant forward converter having one switching element in terms of input feedback noise, the noise component of a direct-current output voltage line, and the like.
FIG. 17 shows an example of configuration of a switching power supply circuit having a voltage resonant converter of the single-ended type. Such a voltage resonant converter is referred to as a multiple resonant converter with a series resonant circuit formed by a leakage inductor L2 on a secondary winding side and a secondary side series resonant capacitor C2 on a secondary side.
In the switching power supply circuit shown in FIG. 17, a rectifying and smoothing circuit formed by a bridge rectifier circuit Di and a smoothing capacitor Ci rectifies and smoothes a commercial alternating-current power AC, and thereby generates a direct-current input voltage Ei as a voltage across the smoothing capacitor Ci. Incidentally, a noise filter that is formed by a set of common mode choke coils CMC and two across capacitors CL and removes common mode noise is provided in the line of the commercial alternating-current power AC.
The direct-current input voltage Ei is input as a direct-current input voltage to the voltage resonant converter. As described above, the voltage resonant converter employs a configuration of the single-ended type with one switching device Q1. The voltage resonant converter in this case is an externally excited converter. The switching device Q1 of a MOSFET is switching-driven by an oscillating and driving circuit 2.
A body diode DD1 of the MOSFET is connected in parallel with the switching device Q1. A primary-side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching device Q1. The primary-side parallel resonant capacitor Cr and a leakage inductor L1 of a primary winding N1 of an isolated converter transformer PIT form a primary side parallel resonant circuit (voltage resonant circuit). This primary side parallel resonant circuit provides a voltage resonant operation as the switching operation of the switching device Q1.
The oscillating and driving circuit 2 applies a gate voltage as a driving signal to the gate of the switching device Q1 to switching-drive the switching device Q1. Thus the switching device Q1 performs switching operation at a switching frequency corresponding to the cycle of the driving signal.
The isolated converter transformer PIT transmits the switching output of the switching device Q1 to the secondary side. As for the structure of the isolated converter transformer PIT, as shown in FIG. 18, for example, the isolated converter transformer PIT has an EE-shaped core formed by combining an E-type core CR1 and an E-type core CR2 of ferrite material with each other. A winding part is divided into a primary side winding part and a secondary side winding part. The primary winding N1 and a secondary winding N2 are wound on a bobbin B covering the central magnetic leg of the EE-shaped core. In addition, a gap G of about 0.8 mm to 1.0 mm is formed in the central magnetic leg of the EE-shaped core of the isolated converter transformer PIT. Thereby a coupling coefficient k=about 0.80 to 0.85 is obtained between the primary side and the secondary side. The coupling coefficient k at this level may be considered to represent loose coupling, and correspondingly a state of saturation is not easily obtained. The value of the coupling coefficient k is a factor in setting leakage inductance (the value of inductance of the leakage inductor L1).
One end of the primary winding N1 of the isolated converter transformer PIT is inserted between the switching element Q1 and the positive electrode terminal of the smoothing capacitor Ci. Thereby, the switching output of the switching element Q1 is transmitted to the primary winding N1. An alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 of the isolated converter transformer PIT.
In this case, a secondary side series resonant capacitor C2 is connected in series with one end of the secondary winding N2. Thus, the leakage inductor L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C2 form a secondary side series resonant circuit (current resonant circuit).
In addition, a voltage doubler half-wave rectifier circuit is formed by connecting rectifier diodes Do1 and Do2 and a smoothing capacitor Co to the secondary side series resonant circuit as shown in the figure. This voltage doubler half-wave rectifier circuit generates a direct-current output voltage Eo having a level corresponding to twice a secondary winding voltage V3 induced in the secondary winding N2 as a voltage across the smoothing capacitor Co. The direct-current output voltage Eo is supplied to a load, and is also input to a control circuit 1 as a detection voltage for constant-voltage control.
The control circuit 1 inputs a detection output obtained by detecting the level of the direct-current output voltage Eo input as the detection voltage to the oscillating and driving circuit 2. The oscillating and driving circuit 2 outputs a driving signal varied in frequency or the like according to the level of the direct-current output voltage Eo indicated by the detection output input to the oscillating and driving circuit 2. The oscillating and driving circuit 2 thereby controls the switching operation of the switching element Q1 so as to make the direct-current output voltage Eo constant at a predetermined level. Thereby control is performed to stabilize the direct-current output voltage Eo.
FIGS. 19A, 19B, and 19C and FIG. 20 show results of experiments on the power supply circuit having the configuration shown in FIG. 17. In conducting the experiments, principal parts of the power supply circuit of FIG. 17 are set as follows.
The isolated converter transformer PIT has an EER-35 as the core, and has a gap length of 1 mm set for the gap of the central magnetic leg. As for the respective numbers T of turns of the primary winding N1 and the secondary winding N2, N1=39 T and N2=23 T. The level of an induced voltage per turn (T) of the secondary winding N2 is set to 3 V/T. The coupling coefficient k of the isolated converter transformer PIT is set to k=0.81.
The capacitance of the primary side parallel resonant capacitor Cr is selected to be Cr=3900 pF (picofarads), and the capacitance of the secondary side series resonant capacitor C2 is selected to be C2=0.1 pF (microfarads). Accordingly, the primary side parallel resonance frequency fo1=230 kHz (kilohertz) of the primary side parallel resonant circuit and the secondary side series resonance frequency fo2=82 kHz of the secondary side series resonant circuit are set. In this case, relative relation between the primary side parallel resonance frequency fo1 and the secondary side series resonance frequency fo2 can be expressed as fo1≈2.8×fo2.
The rated level of the direct-current output voltage Eo is 135 V. Load power handled by the power supply circuit is in a range of maximum load power Pomax=200 W to minimum load power Pomin=0 W.
FIGS. 19A, 19B, and 19C are waveform charts showing the operations of principal parts in the power supply circuit shown in FIG. 17 on the basis of the switching period of the switching element Q1. FIG. 19A shows a switching voltage V1 applied to the switching element Q1, a switching current IQ1, a primary winding current I2, a secondary winding current I3, a rectified current ID1, and a rectified current ID2 at the maximum load power Pomax=200 W. FIG. 19B shows the switching voltage V1, the switching current IQ1, the primary winding current I2, and the secondary winding current I3 at an intermediate load power Po=120 W. FIG. 19C shows the switching voltage V1 and the switching current IQ1 at the minimum load power Pomin=0 W.
The switching voltage V1 is a voltage obtained across the switching element Q1. The switching voltage V1 is at a zero level during a period TON during which the switching element Q1 is on, and forms a sinusoidal resonant pulse waveform during a period TOFF during which the switching element Q1 is off. The resonant pulse waveform of the switching voltage V1 indicates that the operation of the primary side switching converter is of a voltage resonant type.
The switching current IQ1 flows through the switching element Q1 (and the body diode DD1). The switching current IQ1 flows with waveforms shown in the figures during the period TON, and is at a zero level during the period TOFF.
The primary winding current I2 flowing through the primary winding N1 is obtained by combining a current component flowing as the switching current IQ1 during the period TON with a current flowing through the primary side parallel resonant capacitor Cr during the period TOFF. Though shown in only FIG. 19A, in the operation of the secondary side rectifier circuit, the rectified current ID1 and the rectified current ID2 flowing through the rectifier diodes Do1 and Do2 each flow sinusoidally, as shown in the figure. In this case, the resonant operation of the secondary side series resonant circuit appears in the waveform of the rectified current ID1 more dominantly than in the rectified current ID2.
The secondary winding current I3 flowing through the secondary winding N2 has a waveform obtained by combining the rectified current ID1 and the rectified current ID2 with each other. FIG. 20 shows switching frequency fs, the period TON and the period TOFF of the switching element Q1, and AC-to-DC power conversion efficiency (ηAC→DC) of the power supply circuit shown in FIG. 17 with respect to load variation.
First, looking at AC-to-DC power conversion efficiency (ηAC→DC), high efficiencies of 90% or more are obtained over a wide range of load power Po=50 W to 200 W. The present inventor has previously confirmed by experiment that such a characteristic is obtained when a secondary side series resonant circuit is combined with a voltage resonant converter of a single-ended type.
The switching frequency fs, the period TON, and the period TOFF in FIG. 20 indicate the switching operation of the power supply circuit shown in FIG. 17 as a characteristic of constant-voltage control dealing with load variation. In this case, the switching frequency fs is substantially constant with respect to load variation. On the other hand, the period TON and the period TOFF linearly change in manners opposite to each other as shown in FIG. 20. This indicates that the switching operation is controlled by holding the switching frequency (switching period) substantially constant while the direct-current output voltage Eo is varied, and changing a time ratio between the on period and the off period. Such control can be regarded as PWM (Pulse Width Modulation) control that changes the on period and the off period within one cycle. The power supply circuit shown in FIG. 17 stabilizes the direct-current output voltage Eo by this PWM control.
FIG. 21 schematically shows the constant-voltage control characteristic of the power supply circuit shown in FIG. 17 by relation between the switching frequency fs (kHz) and the direct-current output voltage Eo.
The power supply circuit shown in FIG. 17 has the primary side parallel resonant circuit and the secondary side series resonant circuit. Therefore the power supply circuit shown in FIG. 17 has, in a composite manner, two resonant impedance characteristics, that is, a resonant impedance characteristic corresponding to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit and the secondary side series resonance frequency fo2 of the secondary side series resonant circuit. Since the power supply circuit shown in FIG. 17 has the relation fo1≈2.8×fo2, the secondary side series resonance frequency fo2 is lower than the primary side parallel resonance frequency fo1, as shown in FIG. 21.
As for constant-voltage control characteristics with respect to the switching frequency fs under a condition of a constant alternating input voltage VAC, as shown in FIG. 21, constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit are represented by a characteristic curve A and a characteristic curve B, respectively, and constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the secondary side series resonance frequency fo2 of the secondary side parallel resonant circuit are represented by a characteristic curve C and a characteristic curve D, respectively. When constant-voltage control is to be performed at tg, which is a rated level of the direct-current output voltage Eo, under the characteristics shown in FIG. 21, the variable range (necessary control range) of the switching frequency necessary for the constant-voltage control can be represented as a section denoted by Δfs.
The variable range Δfs shown in FIG. 21 extends from the characteristic curve C at the maximum load power Pomax which characteristic curve corresponds to the secondary side series resonance frequency fo2 of the secondary side series resonant circuit to the characteristic curve B at the minimum load power Pomin which characteristic curve corresponds to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit. Crossed between the characteristic curve C at the maximum load power Pomax and the characteristic curve B at the minimum load power Pomin are the characteristic curve D at the minimum load power Pomin which characteristic curve corresponds to the secondary side series resonance frequency fo2 of the secondary side series resonant circuit and the characteristic curve A at the maximum load power Pomax which characteristic curve corresponds to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit.
Thus, as the constant-voltage control operation of the power supply circuit shown in FIG. 17, switching-driving control is performed by PWM control that holds the switching frequency fs substantially fixed, and changes a time ratio (period TON/period TOFF) in one switching period. Incidentally, this is indicated by the fact that the period length of one switching period (TOFF+TON) shown at the times of the maximum load power Pomax=200 W, load power=100 W, and the minimum load power Pomin=0 W in FIGS. 19A, 19B, and 19C is substantially constant, while the widths of the period TOFF and the period TON are changed.
Such an operation is obtained by making a transition between a state in which the resonant impedance (capacitive impedance) at the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit is dominant and a state in which the resonant impedance (inductive impedance) at the secondary side series resonance frequency fo2 of the secondary side parallel resonant circuit is dominant, as resonant impedance characteristics according to load variation in the power supply circuit, in the narrow variable range Δfs of the switching frequency. (See Japanese Patent Laid-Open No. 2000-134925 as Patent Document 1)
The power supply circuit shown in FIG. 17 has the following problems.
The switching current IQ1 at the maximum load power Pomax shown in FIG. 19A is at a zero level until an end of the period TOFF as turn-on timing. When the period TON is reached, the switching current IQ1 operates such that a current of negative polarity first flows through the body diode DD1, and is then inverted to flow from the drain to the source of the switching element Q1. Such an operation indicates that ZVS (Zero Voltage Switching) is performed properly.
On the other hand, the switching current IQ1 at Po=120 W corresponding to an intermediate load which current is shown in FIG. 19B operates so as to flow as noise before the end of the period TOFF as turn-on timing is reached. This operation is an abnormal operation in which ZVS is not properly performed.
That is, it is known that a voltage resonant converter provided with a secondary side series resonant circuit as shown in FIG. 17 causes the abnormal operation in which ZVS is not properly performed at the time of the intermediate load. It is confirmed that the actual power supply circuit shown in FIG. 17 causes such an abnormal operation in a load variation range as a section A shown in FIG. 20, for example.
As described earlier, a voltage resonant converter provided with a secondary side series resonant circuit inherently has, as a tendency thereof, a characteristic of being able to maintain high efficiency in an excellent manner as the load is varied. However, as shown by the switching current IQ1 in FIG. 19B, a considerable peak current flows at the time of turning on the switching element Q1, thereby inviting an increase in switching loss and constituting a factor in lowering power conversion efficiency.
At any rate, an abnormal operation as described above causes a shift in phase-gain characteristics of a constant-voltage control circuit system, for example, thus resulting in a switching operation in a state of abnormal oscillation. Thus, in the present situation, it is generally recognized to be difficult to put the power supply circuit of FIG. 17 to practical use in actuality.